Doherty amplifier and method for operation thereof

ABSTRACT

An amplifier having a Doherty-type architecture and a method for operation thereof are provided. The amplifier comprises a main amplifier path comprising a main amplifier, an auxiliary amplifier path comprising an auxiliary amplifier, and an signal preparation unit configured to develop a main amplifier input signal for the main amplifier path and an auxiliary amplifier input signal for the auxiliary amplifier path based on an amplifier input that is to be amplified and a transition threshold associated with the amplifier input. By driving the main and auxiliary amplifiers as a function of the transition threshold, the gain of the Doherty-type amplifier may be increased.

RELATED APPLICATION

The present patent application is a continuation of U.S. patentapplication Ser. No. 12/482,110, filed Jun. 10, 2009, now U.S. Pat. No.8,022,768 and claims the benefit of U.S. Provisional Patent ApplicationNo. 61/139,244 filed Dec. 19, 2008, the entire contents of which areincorporated herein by reference.

FIELD OF THE INVENTION

The present invention relates to signal power amplification, and morespecifically to Doherty-type power amplifiers.

BACKGROUND

Wireless devices use high frequency, generally referred to as RadioFrequency (RF), signals to transmit information through free space. Forexample, cell phones use amplified RF to transmit voice data to basestations, which allow signals to be relayed to communications networks.Other existing wireless communication devices include Bluetooth, HomeRFand WLAN. Note that the term “Radio Frequencies” is used herein to referto any frequencies suitable for wireless communication, including UltraHigh Frequency (UHF), Very High Frequency (VHF), Microwave Frequency,etc.

In a conventional wireless device, the power amplifier generallyconsumes most of the power of the overall wireless system. For systemsthat run on batteries, a power amplifier with a low efficiency(P_(out)/P_(supply)) results in a reduced communication time for a givenbattery life. A decrease in efficiency of the power amplifier typicallyresults in increased power usage and heat removal requirements, whichmay increase the equipment and operating costs of the overall system.

For this reason, much effort has been expended on increasing theefficiency of RF power amplifiers.

A class AB amplifier, that is an amplifier with amplifying device(s)biased in class AB is typically considered the standard against whichother amplifier architectures are compared in terms of gain andefficiency.

One type of amplifier that may increase power amplifier efficiency is aDoherty-type power amplifier. A common Doherty-type power amplifierdesign includes a main amplifier and an auxiliary amplifier. The mainamplifier is operated to maintain optimal efficiency up to a certainpower level and allows the auxiliary amplifier to operate above thatlevel. When the power amplifier is operated at a high output powerlevel, the gain of the main amplifier will be heavily compressed suchthat non-linearities are introduced into the amplified signal.

In a conventional Doherty-type amplifier, a signal preparation unit,often implemented with a simple power splitting structure, is used todivide an amplifier input signal along main and auxiliary amplificationpaths to the main and auxiliary amplifiers, respectively, foramplification.

FIG. 1 is a block diagram of a conventional Doherty-type amplificationunit 100. As shown in FIG. 1, Doherty-type amplification unit 100comprises an input signal line 106, a main amplifier 102, an auxiliaryamplifier 104, a signal preparation unit 110, a main amplifier impedancetransformer 112, and an output signal line 108. An input signal ispassed into input signal line 106 and into signal preparation unit 110.Signal preparation unit 110 transmits the input signal from input signalline 106 into main amplifier 102, and signal preparation unit 110 phaseshifts the input signal from input signal line 106 and transmits thephase shifted signal to auxiliary amplifier 104. A combining structure114, which includes the main amplifier impedance transformer 112 in themain amplifier signal path at the output of the main amplifier 102,receives output from the main amplifier 102 and output from theauxiliary amplifier 104 and combines them to form an output signal thatis transmitted to signal output line 108.

In some conventional Doherty-type amplifiers, the phase shift introducedby signal preparation unit 110 is corrected in main amplifier impedancetransformer 112, so that the signal exiting main amplifier impedancetransformer 112 is in phase with the signal that exits auxiliaryamplifier 104.

In many conventional Doherty-type amplifiers, a matching structure (notshown in FIG. 1) is used to match the output impedance of the Main andAuxiliary amplifiers to the input impedance of the device that is beingdriven by the output of the Doherty-type amplifier, typically an antennastructure.

Although Doherty-type amplifiers are often superior in terms ofefficiency over Class AB amplifiers, they are often plagued by reducedgain levels. Typical Doherty-type amplifier designs (Classical,Asymmetrical and Enhanced asymmetrical) generally incorporate a powerdivider that either separates the input signal power equally, or in apre-set ratio, to the Main and Auxiliary amplifier paths. However, theportion of the input signal that is directed to the Auxiliary amplifierpath is typically used to self-bias the Auxiliary amplifier in back-offand therefore does not develop measurable power at the output of theAuxiliary amplifier since the Auxiliary amplifier is typically offduring back-off operation thereby reducing the overall gain of theconventional Doherty-type amplifier. In other words, the portion of theinput signal that is directed to the Auxiliary amplifier in aconventional Doherty-type amplifier is not amplified during back-offoperation thereby lowering the overall gain.

FIG. 2 is a plot of gain and efficiency versus output power for aconventional AB amplifier at 200, 202 respectively and a plot of gainand efficiency versus output power for a conventional Doherty-typeamplifier at 206, 204 respectively. It can be seen from FIG. 2 thatefficiency 200 of the class AB amplifier is much poorer than theefficiency 204 of the conventional Doherty-type amplifier. However, thegain 202 of the class AB amplifier is almost 4 dB higher than the gain206 of the conventional Doherty-type amplifier.

In order to increase the gain of a conventional Doherty-type amplifiers,larger drive circuitry, i.e., larger Main and Auxiliary path drivers (oramplifiers prior to the Signal Preparation Unit, 110) that consume morepower, are typically used, which quickly reduces the efficiencyimprovements achieved through use of the Doherty architecture.

In common Doherty-type amplifiers, the main and auxiliary amplifiers arecomposed of the same type of amplifiers with the same poweramplification rating. These Doherty-type amplifiers develop anefficiency peak 6 dB back of full power which in theory will be equal inmagnitude to the maximum efficiency of the system.

In asymmetrical Doherty-type amplifiers, the main and auxiliaryamplifiers are implemented with devices of unequal size in terms ofpower. Accordingly, traditional Doherty characteristics, specificallyefficiency versus output power, can be modified in order to adjust thelocation of the peak efficiency in back-off.

A further advance over asymmetrical Doherty-type amplifiers is providedin an enhanced asymmetrical Doherty-type amplifier, as described in U.S.Patent Application Publication No. US 2008/0088369, published Apr. 17,2008, which is assigned to the Assignee of this application, and ishereby incorporated by reference in its entirety. An enhancedasymmetrical Doherty-type amplifier utilizes an asymmetricalDoherty-type amplifier structure with Main and Auxiliary amplifierdevices of different semiconductor technologies to take advantage of theperformance characteristics, for example, linearity and power handlingcharacteristics, of the different semiconductor technologies.

With enhanced asymmetrical Doherty-type amplifiers, relatively highefficiency can be tailored to a given output power transfer functionwith different sizes, in terms of power, for the Main and Auxiliaryamplifiers, as well as the use of different semiconductor technologiesto implement the Main and Auxiliary amplifiers. In some implementations,various classes of device biasing are used to bias the devices used toimplement the Main and Auxiliary amplifiers. Examples of potentialdevice biasing classes include, but are not limited to, Class A, AB, B,C, D and H.

An amplifier device biased in class A conducts current at all times,Class B amplifiers are designed to amplify half of an input wave signal,and Class AB is intended to refer to the Class of amplifier whichcombines the Class A and Class B amplifier. As a result of the Class Bproperties, Class AB amplifiers are operated in a non-linear region thatis only linear over half the wave form. Class C amplifiers are biasedwell beyond cut-off, so that current, and consequently the input signal,is amplified less than one half the duration of any given period. TheClass C design provides higher power-efficiency than Class B operationbut with the penalty of higher input-to-output nonlinearity.

Furthermore, these types of amplifiers are high in memory, i.e.,previous operating states effect the current state, and therefore aredifficult to “correct” or “linearize”, due to their complex gain andphase profiles.

Asymmetrical Doherty-type amplifiers are even more difficult tolinearize as the typically smaller Main amplifier is pushed deeper intocompression before the Auxiliary amplifier turns on to handle the higherinput power levels.

Enhanced asymmetrical Doherty-type amplifiers are again more difficultto linearize due to the asymmetrical transfer functions, specifically interms of gain and phase, which the mixed semiconductor devicesintroduce.

Conventional pre-distortion algorithms typically cannot providesufficient correction of asymmetrical and/or enhanced asymmetricalDoherty-type amplifiers such that they comply with transmissionstandards such as CDMA, UMTS, HSPA, OFDM, WiMAX, LTE and multicarrierGSM. Consequently, unlinearized enhanced asymmetrical Doherty amplifiersare generally unsuitable for linear modulation systems.

In addition to the gain and linearization issues, conventionalDoherty-type amplifiers typically only operate over a narrow frequencyband, which is typically limited by the Doherty combining and matchingstructures utilized to combine the outputs of the main and auxiliaryamplifier paths and match the outputs of the main and auxiliaryamplifiers to a load impedance. Accordingly, conventional Doherty-typeamplifiers are typically limited to narrow band applications.

SUMMARY OF THE INVENTION

According to one broad aspect of the present invention, there isprovided an amplifier arrangement for amplifying an amplifierarrangement input, the amplifier comprising: a main amplifier pathcomprising a main amplifier; an auxiliary amplifier path comprising anauxiliary amplifier; a combining structure configured to combine outputsof the main and auxiliary amplifiers; and a signal preparation unitconfigured to develop a main amplifier input signal for the mainamplifier path and an auxiliary amplifier input signal for the auxiliaryamplifier path as a function of the amplifier arrangement input and atransition threshold associated with the amplifier arrangement input.

In some embodiments, the amplifier arrangement further comprises a biascontroller configured to bias the auxiliary amplifier substantially atbut below a turn-on voltage of the auxiliary amplifier.

In some embodiments, the signal preparation unit is configured to:develop the main amplifier input signal with substantially all of theamplifier arrangement input for voltages of the amplifier arrangementinput that are below the transition threshold; and redirect at leastsome portion of the amplifier arrangement input to develop the auxiliaryamplifier input signal for voltages of the amplifier arrangement inputthat are above the transition threshold.

In some embodiments, the transition threshold is substantially equal tothe 1 dB compression point (P1 dB) of the main amplifier.

In some embodiments, the signal preparation unit is configured toasymmetrically divide the amplifier arrangement input above thetransition threshold between the main amplifier input signal and theauxiliary amplifier input signal.

In some embodiments, the signal preparation unit is configured toasymmetrically divide the amplifier arrangement input above thetransition threshold based on a ratio between maximum power ratings ofthe main amplifier and the auxiliary amplifier.

In some embodiments, the amplifier arrangement further comprises: aplurality of auxiliary amplifier paths each comprising a respectiveauxiliary amplifier, inclusive of the first recited auxiliary amplifierpath comprising the first recited auxiliary amplifier, wherein thesignal preparation unit is configured to develop the main amplifierinput signal for the main amplifier path and a respective auxiliaryamplifier input signal for each one of the plurality of auxiliaryamplifier paths, inclusive of the first recited auxiliary amplifierinput signal for the first recited auxiliary amplifier path, based on aplurality of transition thresholds inclusive of the first recitedtransition threshold.

In some embodiments: electrical length of the auxiliary amplifier pathafter the auxiliary amplifier is substantially equal to N×180°, where Nis an integer; and electrical length of the main amplifier path afterthe main amplifier is 90° longer than the electrical length of theauxiliary amplifier path after the auxiliary amplifier.

In some embodiments, the amplifier arrangement further comprises amatching structure configured to substantially match the outputs of themain and auxiliary amplifiers to a desired load impedance.

In some embodiments: the combining structure forms part of the main andauxiliary amplifier paths and comprises a quarter wave transformerconfigured to provide the additional 90° electrical length in the mainamplifier path after the main amplifier relative to the auxiliaryamplifier path after the auxiliary amplifier; and the quarter wavetransformer comprises any one of: a microstrip; and a shorted stub.

In some embodiments, the main amplifier and the auxiliary amplifier areasymmetrically sized in terms of maximum power ratings.

In some embodiments, the main amplifier and the auxiliary amplifier arefabricated from different semiconductor materials.

In some embodiments, the main amplifier and the auxiliary amplifier areprovided in a common package to minimize the electrical lengths of themain and auxiliary amplifier paths.

In some embodiments, the combining structure is provided in the commonpackage to minimize the electrical length of the combining structure.

According to another broad aspect of the present invention, there isprovided a method for controlling a Doherty amplifier comprising a mainamplifier and an auxiliary amplifier, the method comprising: developinga main amplifier input signal for the main amplifier and an auxiliaryamplifier input signal for the auxiliary amplifier as a function of anamplifier input that is to be amplified and a transition thresholdassociated with the amplifier input.

In some embodiments, the method further comprises biasing the auxiliaryamplifier substantially at but below a turn-on voltage of the auxiliaryamplifier.

In some embodiments, developing the main amplifier input signal and theauxiliary amplifier input signal comprises: for voltages of theamplifier input that are below the transition threshold, developing themain amplifier input signal with substantially all of the amplifierinput; and for voltages of the amplifier input that are above thetransition threshold, redirecting at least some portion of the amplifierinput to develop the at least one auxiliary amplifier input signal.

In some embodiments, the transition threshold is substantially equal tothe 1 dB compression point (P1 dB) of the main amplifier.

In some embodiments, redirecting at least some portion of the amplifierinput comprises: asymmetrically dividing the amplifier input above thetransition threshold between the main amplifier input signal and theauxiliary amplifier input signal.

In some embodiments, asymmetrically dividing the amplifier input abovethe transition threshold comprises dividing the amplifier input based ona ratio between maximum power ratings of the main amplifier and theauxiliary amplifier.

Other aspects and features of the present invention will becomeapparent, to those ordinarily skilled in the art, upon review of thefollowing description of the specific embodiments of the invention.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the invention will now be described in greater detailwith reference to the accompanying drawings, in which:

FIG. 1 is a block diagram of a known Doherty-type amplification unit;

FIG. 2 is a plot of gain and efficiency versus output power for aconventional class AB amplifier and a conventional Doherty-typeamplifier;

FIG. 3 is a block diagram of a Doherty-type amplifier according to anembodiment of the invention;

FIG. 4A is a plot of simulated gain and efficiency for a Doherty-typeamplifier with an auxiliary amplifier biased in class C according to anembodiment of the present invention;

FIG. 4B is a plot of simulated gain and efficiency for a Doherty-typeamplifier with an auxiliary amplifier biased in class AB according to anembodiment of the present invention;

FIG. 4C is a plot of simulated gain and efficiency for a Doherty-typeamplifier with an auxiliary amplifier biased in class B according to anembodiment of the present invention;

FIG. 5A is a plot of simulated gain and efficiency for a Doherty-typeamplifier with a transition threshold set below the 1 dB compressionpoint of the main amplifier according to an embodiment of the presentinvention;

FIG. 5B is a plot of simulated gain and efficiency for a Doherty-typeamplifier with a transition threshold set above the 1 dB compressionpoint of the main amplifier according to an embodiment of the presentinvention;

FIG. 5C is a plot of simulated gain and efficiency for a Doherty-typeamplifier with a transition threshold set substantially at the 1 dBcompression point of the main amplifier according to an embodiment ofthe present invention;

FIGS. 6A and 6B are schematic diagrams of a Doherty-type amplifier withthe auxiliary amplifier in an “off” state and in an “on” state,respectively;

FIG. 6C is a plot of simulated gain and efficiency of an amplifierbiased in class AB and presented with two different load impedancescorresponding to the load impedances seen by the main amplifier of theDoherty-type amplifier shown in FIGS. 6A and 6B, with the auxiliaryamplifier in the “off” state and the “on” state, respectively;

FIG. 7A is a plot of simulated gain and efficiency for a Doherty-typeamplifier with all subsequent input power above a transition thresholdvoltage directed to the auxiliary amplifier according to an embodimentof the present invention;

FIG. 7B is a plot of simulated gain and efficiency for a Doherty-typeamplifier with all subsequent input power above a transition thresholdvoltage split equally between main amplifier and the auxiliary amplifieraccording to an embodiment of the present invention;

FIG. 7C is a plot of simulated gain and efficiency for a Doherty-typeamplifier with all subsequent input power above a transition thresholdvoltage split unequally between main amplifier and the auxiliaryamplifier according to an embodiment of the present invention;

FIG. 8A is a plot of transfer functions relating main and auxiliaryamplifier input signal voltages to input power resulting in equal mainand auxiliary amplifier input signals in a known asymmetric Doherty-typeamplifier;

FIG. 8B is a plot of the simulated gain and efficiency for variousauxiliary bias voltage in a Doherty-type amplifier driven with equalmain and auxiliary amplifier input signals generated according to thetransfer functions shown in FIG. 8A;

FIG. 9A is a plot of transfer functions relating main and auxiliaryamplifier input signal voltages to input power resulting in unequal mainand auxiliary amplifier input signals in a known asymmetric Doherty-typeamplifier;

FIG. 9B is a plot of the simulated gain and efficiency for variousauxiliary bias voltages in a Doherty-type amplifier driven with unequalmain and auxiliary amplifier input signals generated according to thetransfer functions shown in FIG. 9A;

FIG. 10A is a plot of transfer functions relating main and auxiliaryamplifier input signals to input power wherein all input power below atransition threshold voltage is directed to the main amplifier inputsignal and thereafter asymmetric post-transition power division is usedaccording to an embodiment of the present invention;

FIG. 10B is a plot of the simulated gain and efficiency for variousauxiliary bias voltages in a Doherty-type amplifier driven with the mainand auxiliary amplifier input signals shown in FIG. 10A;

FIG. 11A is a schematic diagram of an idealized Doherty combiningstructure;

FIG. 11B is a Smith chart plot of the simulated s-parameter s11 for theidealized Doherty combining structure of FIG. 11A;

FIG. 12A is a schematic diagram of an improvised implementation of theidealized Doherty combining structure of FIG. 11A;

FIG. 12B is a Smith chart plot of the simulated s-parameter s11 for theimprovised Doherty combining structure of FIG. 12A;

FIGS. 13A and 13B are schematic diagrams of Doherty-type amplifiers withdifferent Doherty combining structures in accordance with embodiments ofthe present invention;

FIG. 14A is a plot of the simulated s-parameter s11 for the Doherty-typeamplifiers shown in FIGS. 13A and 13B;

FIG. 14B is a plot of the simulated phase of the s-parameter s21 for theDoherty-type amplifiers shown in FIGS. 13A and 13B;

FIG. 15A is a Smith chart plot of the measured s-parameter s22 for aDoherty-type amplifier in accordance with an embodiment of the presentinvention, with the measurement equipment calibrated at a firstreference plane of the device under test;

FIG. 15B is a Smith chart plot of the measured s-parameter s22 for aDoherty-type amplifier in accordance with an embodiment of the presentinvention, with the measurement equipment calibrated at a secondreference plane of the device under test;

FIG. 15C is a Smith chart plot of the measured s-parameter s22 for aDoherty-type amplifier with the electrical length of the Doherty outputstructure optimized across frequency in accordance with an embodiment ofthe present invention;

FIG. 16 is a schematic diagram of the Doherty-type amplifier on whichthe s-parameter s22 measurements shown in FIGS. 15A to 15C were made;

FIG. 17A is a plot of measured gain versus output power across frequencyfor the Doherty-type amplifier shown in FIG. 16, both with and withoutthe optimization of the Doherty output structure, according to anembodiment of the invention;

FIG. 17B is a plot of measured efficiency versus output power acrossfrequency for the Doherty-type amplifier shown in FIG. 16, both with andwithout the optimization of the Doherty output structure, according toan embodiment of the invention; and

FIG. 17C is a plot of the measured phase introduced by the Doherty-typeamplifier shown in FIG. 16 for various frequencies, both with andwithout the optimization of the Doherty output structure, according toan embodiment of the invention.

DETAILED DESCRIPTION

In the following detailed description of sample embodiments, referenceis made to the accompanying drawings, which form a part hereof, and inwhich is shown by way of illustration specific sample embodiments inwhich the present invention may be practised. These embodiments aredescribed in sufficient detail to enable those skilled in the art topractice the invention, and it is to be understood that otherembodiments may be utilized and that logical, mechanical, electrical,and other changes may be made without departing from the scope of theinvention. The following detailed description is, therefore, not to betaken in a limiting sense, and the scope is defined by the appendedclaims.

Various architectures and methods relating to amplifiers having aDoherty-type architecture, that is an amplifier architecture featuring amain amplification path and one or more auxiliary amplification paths inparallel with the main amplification path, are provided.

Increased gain for Doherty-type amplifier architectures is potentiallyachieved in some embodiments of the present invention by controlling theinput drive signals provided to the main and auxiliary amplifier pathssuch that substantially all of the power in an input signal that is tobe amplified is directed to the main amplifier path up to a transitionthreshold, after which a post-transition threshold power divisionstrategy is employed to control division of the input signal between themain and auxiliary amplifier paths. A bias value for the auxiliaryamplifier(s) in the auxiliary amplifier path may be selected to controlwhen the auxiliary amplifier(s) turn-on in relation to the threshold.

Details of how input signal preparation, auxiliary amplifier biasing,transition threshold selection and post-transition threshold powerdivision may be determined are provided below by way of example withreference to specific embodiments. The specific embodiments discussedbelow are provided for illustrative purposes only and should not beconstrued as limiting as to the broader aspects of the presentdisclosure.

FIG. 3 is a block diagram of a Doherty-type amplifier arrangement 300 inaccordance with an embodiment of the present invention. The Doherty-typeamplifier arrangement 300 includes a signal preparation unit 306, mainand auxiliary amplifier paths that include a main amplifier 308 and anauxiliary amplifier 310, respectively and an output signal combiner 312that is operatively connected to outputs of the main and auxiliaryamplifiers 308, 310.

The main and auxiliary amplifiers 308, 310 each have a bias controlinput through which a bias voltage Vbias_main 318 and Vbias_aux 320,respectively can be set. Details of the selection of the bias voltagesVbias_main 318 and Vbias_aux 320 that might be used for some embodimentsof the present invention are discussed below with reference to FIG. 4.

In some embodiments, the amplifier arrangement 300 includes a biascontroller (not shown) that controls the biasing of the main andauxiliary amplifiers.

In operation, the signal preparation unit 306 develops main andauxiliary amplifier path input signals 322, 324 that it drives to themain and auxiliary amplifiers 308, 310 respectively based on anamplifier input signal 302 received at an input of the signalpreparation unit and a transition threshold. Details of how the inputsignal preparation unit 306 develops main and auxiliary amplifier pathinput signals 322, 324 based on the input signal 302 and a transitionthreshold are discussed in detail below with reference to FIGS. 5 to 10.

The output signal combiner 312 combines outputs of the main andauxiliary amplifiers 308, 310 resulting in an amplified output signal304.

In some embodiments, the output signal combiner 312 includes a combiningblock 314 that combines the outputs of main and auxiliary amplifiers308, 310 and a matching block 316 that is intended to substantiallymatch the combined output impedance of the combining block to an inputimpedance of another element. In the context of a wireless transmitter,the next element in the transmitter following a power amplifier, such asthe Doherty-type amplifier arrangement 300 shown in FIG. 3, is typicallyan isolator, filter, duplexer or antenna. Accordingly, in someembodiments, the matching block 316 is intended to substantially matchan output impedance of the combining block 314 with the input impedanceof one of those elements (not shown). While the output signal combiner312 shown in FIG. 3 includes separate combining and matching blocks 314and 316, in some embodiments some elements, such as a quarter wavetransformer in the output path of the main amplifier 308, may serve asboth a combining and a matching element.

In some embodiments, the main amplifier 308 and/or the auxiliaryamplifier 310 include their own matching structures so that thesemiconductor structures used to realize the main and auxiliaryamplifiers are presented with desired target impedances.

While the Doherty-type amplifier 300 shown in FIG. 3 only includes asingle auxiliary amplifier path with a single auxiliary amplifier, moregenerally a Doherty-type amplifier arrangement in accordance with anembodiment of the present invention may include any number of auxiliaryamplifier paths each having a respective auxiliary amplifier.

For embodiments with multiple auxiliary amplifier paths, the signalpreparation unit may develop the amplifier input signals for each of theamplifier paths based on multiple transition thresholds utilizingasymmetric post-transition threshold power division as described herein,such that successive auxiliary amplifier paths receive some portion ofthe input signal power as the input signal power transitions throughtheir corresponding transition thresholds. The use of transitionthresholds in developing main and auxiliary amplifier input signals andpost-transition threshold power division strategies are discussed infurther detail below with reference to single auxiliary amplifier pathembodiments. However, it should be understood that those features areextendable to embodiments that include more than one auxiliary amplifierpath.

Furthermore, it should be understood that although the main amplifier308 and the auxiliary amplifier 310 are shown as individual amplifierelements in FIG. 3, more generally the main amplifier and the auxiliaryamplifier may include any number of amplifiers arranged in successivegain stages.

Selection of the biasing class of the auxiliary amplifier 310 will nowbe described with reference to FIGS. 4A to 4C, which illustrate plots ofoverall gain and efficiency versus output signal power of a Doherty-typeamplifier arrangement in accordance with an embodiment of the presentinvention for three different auxiliary amplifier bias settings.

FIG. 4A shows exemplary gain 400 and efficiency 402 curves for aDoherty-type amplifier in accordance with an embodiment of the presentinvention in which the auxiliary amplifier is biased in class C. Asillustrated in region 404 of the FIG. 4, class C biasing of theauxiliary amplifier causes a lag before the auxiliary amplifier beginsto add to the output power. This is because in class C biasing, theauxiliary amplifier is biased such that it is turned-off until theauxiliary amplifier path input signal 324 is strong enough to turn onthe auxiliary amplifier, which causes a drop-off in the gain as theportion of the power of the input signal 302 that is diverted to developthe auxiliary amplifier input signal 324 is not amplified until theauxiliary amplifier is turned on. Biasing the auxiliary amplifier inclass C typically results in a system that is heavily compressed.Furthermore, when biased in Class C the auxiliary amplifier 310 demandshigher drive levels, as compared to biasing in class AB or class B,before the auxiliary amplifier contributes to the overall gain of theDoherty amplifier.

FIG. 4B shows exemplary gain 410 and efficiency 412 curves for aDoherty-type amplifier in accordance with an embodiment of the presentinvention in which the auxiliary amplifier is biased in class AB. Whenbiased in class AB, the auxiliary amplifier provides amplified output assoon as any auxiliary amplifier input signal 324 is provided to it. Thiscauses discontinuities in the gain and efficiency curves 410, 412, whichcan be seen in regions 414 and 416 of FIG. 4B. These discontinuities canmake it difficult for pre-distortion algorithms to linearizeDoherty-type amplifiers with auxiliary amplifiers biased in class AB.However, biasing the auxiliary amplifier in class AB can potentiallyincrease the efficiency at higher output power levels, as shown in FIG.4B

FIG. 4C shows exemplary gain 420 and efficiency 422 curves for aDoherty-type amplifier in accordance with an embodiment of the presentinvention in which the auxiliary amplifier is biased in class B. InClass B, the auxiliary amplifier is biased substantially at or near itsturn-on voltage, so that the lag present in class C biasing issubstantially eliminated, as the auxiliary amplifier substantiallyimmediately provides unity gain (0 dB). Additionally, the discontinuityin the gain and efficiency with class AB biasing is substantiallyeliminated with class B biasing, as an auxiliary amplifier biased inclass B transitions from unity gain (0 dB) to its maximum gain in acontinuous manner. This can potentially make a Doherty-type amplifierwith an auxiliary amplifier biased in class B easier to linearize thanone in which the auxiliary amplifier is biased in class AB.

With further reference to FIG. 3, the biasing class of the auxiliaryamplifier, which is controlled by the auxiliary amplifier bias controlvoltage Vbias_aux 320, determines how the auxiliary amplifier respondsto the auxiliary amplifier input signal 324 provided by the input signalsplitter 306. The operation of the signal preparation unit 306 inproviding the main and auxiliary amplifier input signals 322 and 324,respectively, from the input signal 302 will now be described, by way ofexample, with reference to FIGS. 6 to 10.

Many conventional Doherty-type amplifiers include a signal preparationunit that uses symmetrical power division to develop input signals formain and auxiliary amplifier. For example, referring to FIG. 1 again,the conventional signal preparation unit 110 may symmetrically dividethe power of the input signal 106 between the main and auxiliaryamplifier paths to develop the main and auxiliary amplifier inputsignals to the main and auxiliary amplifiers 102 and 104 such that themain amplifier 102 and the auxiliary amplifier 104 are driven equally.

FIG. 8A is a plot of transfer functions 800 and 802 relating main andauxiliary amplifier input signal voltages, respectively, to the inputpower of a Doherty-type amplifier. The transfer functions 800 and 802result in equal/symmetric main and auxiliary amplifier input signals.FIG. 8B is a plot of gain curves 804, 806, 808, 810 and 812 andefficiency curves 814, 816, 818, 820 and 822 versus amplifier outputpower for a Doherty-type amplifier with symmetric main and auxiliaryamplifier input signals generated according to the transfer functions800 and 802 illustrated in FIG. 8A. Each gain and efficiency curve pair804 and 814, 806 and 816, 808 and 818, 810 and 820, and 812 and 822,corresponds to a different auxiliary amplifier bias voltage.

As can be seen in FIG. 8B, when symmetrical power division is used todevelop main and auxiliary amplifier input signals, the efficiencycurves 820 and 822 for lower auxiliary amplifier bias voltages (moretowards class C biasing of the auxiliary amplifier) are lower than theefficiency curves 814, 816 and 818 for higher auxiliary amplifier biasvoltages (more towards class B biasing of the auxiliary amplifier).

In some conventional Doherty-type amplifiers, asymmetric power divisionmay be used to develop the main and auxiliary amplifier input signals byasymmetrically dividing the power of an input signal between the mainand auxiliary amplifier paths. FIG. 9A is a plot of transfer functions900 and 902 relating main and auxiliary amplifier input signal voltages,respectively, to the input signal power of a Doherty-type amplifier,which result in unequal/asymmetric main and auxiliary amplifier inputsignals. The asymmetric power division implemented by the transferfunctions 900 and 902 of FIG. 9A are such that the auxiliary amplifierinput signal receives a greater percentage of the power of the inputsignal than the main amplifier input signal for all input signal powerlevels.

FIG. 9B is a plot of gain curves 904, 906, 908, 910 and 912 andefficiency curves 914, 916, 918, 920 and 922 versus amplifier outputpower for a conventional Doherty-type amplifier with asymmetric inputsignal power division according to the transfer functions 900 and 902illustrated in FIG. 9A. As can be seen in FIG. 9B, for conventionalasymmetrical power division of an input signal between main andauxiliary amplifiers, the efficiency curves 920 and 922 for lowerauxiliary amplifier bias voltages are lower than the efficiency curves914, 916 and 918 for higher auxiliary amplifier bias voltages, which,like the symmetrical power division results illustrated in FIG. 8B,indicates that the biasing of the auxiliary amplifier impacts theefficiency of a Doherty-type amplifier operating with asymmetric orsymmetric power division. However, it is noted that although the peakgain (approx. 10 dB) of the gain curves 904, 906, 908, 910 and 912 forasymmetric power division is lower than the peak gain (approx. 12 dB) ofthe gain curves 804, 806, 808, 810 and 812 for symmetric power division,the gain curves 904, 906, 908, 910 and 912 for asymmetric power divisionappear to be more linear than the corresponding gain curves 804, 806,808, 810 and 812 for symmetric power division.

Referring again to FIG. 3, in contrast to the conventional powerdivision strategies shown in FIGS. 8A and 9A, the signal preparationunit 306 shown in FIG. 3 develops the main and auxiliary amplifier inputsignals 322 and 324, respectively, based on the input signal 302 and adetermination of whether or not the input signal 302 has exceeded atransition threshold.

More specifically, in some embodiments the signal preparation unit 306develops the main and auxiliary amplifier input signals 322 and 324,respectively, by developing the main amplifier input signal 322 withsubstantially all of the power of the input signal 302 up to atransition threshold, after which some or all of the additional inputsignal 302 above the threshold level is used to develop the auxiliaryamplifier input signal 324. Effectively, in these embodiments the signalpreparation unit can be thought of as a switch that, for input signallevels below the threshold, diverts substantially all of the inputsignal to the main amplifier 308 and at the transition threshold inputsignal level begins to divert at least some portion of the input signalto the auxiliary amplifier 310.

FIG. 10A is a plot of transfer functions 1000 and 1002 relating main andauxiliary amplifier input signal voltages, respectively, to input signalpower, that include a transition threshold and utilize asymmetricpost-transition threshold power division. In FIG. 10A, for input signalpower levels below a threshold power level 1003, the main amplifierinput signal is developed with substantially all of the input signalpower and after the threshold power level 1003 the additional inputsignal power is asymmetrically divided between the main amplifier inputsignal and the auxiliary amplifier input signal such that after thetransition threshold power level 1003 the auxiliary amplifier receives agreater percentage of the additional input signal power. This causes theauxiliary amplifier input signal to increase relative to the mainamplifier input signal after the transition threshold power level 1003,which causes the plot of main amplifier input signal 1000 and the plotof the auxiliary amplifier input signal 1002 to intersect at an inputsignal power 1005, after which the auxiliary amplifier input signal islarger than the main amplifier input signal.

In some embodiments, asymmetric post-transition power division isperformed such that the input signal power 1005 at which the mainamplifier input signal and the auxiliary amplifier input signal areequalized occurs at the 1 dB compression point of the main amplifier.

FIG. 10B is a plot of gain curves 1004, 1006, 1008, 1010 and 1012 andefficiency curves 1014, 1016, 1018, 1020 and 1022 versus amplifieroutput power for a Doherty-type amplifier with asymmetricpost-transition threshold power division according to the transferfunctions 1000 and 1002 illustrated in FIG. 10A. As can be seen in FIG.10B, for asymmetric post-transition threshold power division, theefficiency curves 1004, 1006, 1008, 1010 and 1012 are virtuallyidentical, showing little change in efficiency for different auxiliarybias voltages. Furthermore, this approach results in the gain curves1014, 1016, 1018, 1020 and 1022 having higher peak gain (approx. 15 dB)compared to the corresponding gain curves illustrated in FIGS. 8B and9B. In FIG. 10B, the gain curves 1012, 1010, 1008, 1006 and 1004correspond to increasing levels of auxiliary bias voltage. It is clearfrom FIG. 10B that the higher the auxiliary bias voltage, the morelinear the gain response of the overall Doherty-type amplifier, forexample, the gain curve 1004 is more linear than the gain curve 1012.

Post threshold power division between the main and auxiliary amplifiersis discussed in further detail below with reference to FIGS. 5 to 7.

For a Doherty-type amplifier that utilizes post-transition thresholdpower division as described herein, selection of the input signaltransition threshold relative to the 1 dB compression input voltage, P1dB, of the main amplifier can potentially impact the linearity andefficiency of the Doherty-type amplifier. FIGS. 5A to 5C are plots ofgain and efficiency curves versus output signal power for threedifferent transition thresholds relative to the P1 dB input voltage ofthe main amplifier of a Doherty-type amplifier in accordance with anembodiment of the present invention in which asymmetric post-transitionthreshold power division directs a majority of the input signal powerabove the transition threshold to the auxiliary amplifier (similar tothe asymmetric post-transition threshold power division illustrated inFIG. 10A).

FIG. 5A is a plot of a gain curve 500 and an efficiency curve 502 of aDoherty-type amplifier operating with asymmetric post-transitionthreshold power division similar to the asymmetric post-transitionthreshold power division illustrated in FIG. 10A, in which thetransition threshold is selected to be less than the P1 dB input voltageof the main amplifier of the Doherty-type amplifier. As can be seen fromFIG. 5A, when the transition threshold is set to a value that is lessthan the P1 dB input voltage of the main amplifier and most of the inputsignal power is directed to the auxiliary amplifier above the transitionthreshold, the drive of the main amplifier towards compression isterminated prematurely, which means that maximum efficiency in back-offis not achieved. In this embodiment, the drop off, generally indicatedat 504, in the gain curve 504 occurs at the transition threshold beforethe P1 dB of the main amplifier is reached, so that the increasingefficiency that comes with driving the main amplifier towardscompression levels off briefly at approximately 35% efficiency,generally indicated at 506, and does not approach 60% efficiency untilthe output power reaches approximately 55 dBm.

FIG. 5B is a plot of a gain curve 510 and an efficiency curve 512 of aDoherty-type amplifier operating with asymmetric post-transitionthreshold power division similar to the asymmetric post-transitionthreshold power division illustrated in FIG. 10A, in which thetransition threshold is selected to be greater than the P1 dB inputvoltage of the main amplifier of the Doherty-type amplifier. As can beseen from FIG. 5B, when the transition threshold is set to a value thatis greater than the P1 dB input voltage of the main amplifier, the mainamplifier of the Doherty-type amplifier is over-driven such that itexperiences excessive compression, generally indicated by the drop offin the gain curve 510 at 514. Essentially, when the transition level isset above the P1 dB compression voltage of the main amplifier, the gainprovided by the main amplifier is heavily compressed before any gaincontribution from the auxiliary amplifier is realized. Furthermore,because of the heavy compression, i.e., non-linear, performance, of theover-driven main amplifier above its P1 dB input voltage, pre-distortionalgorithms will generally have difficulties linearizing a Doherty-typeamplifier operated in this mode. However, it should be noted thatbecause the transition threshold is above the P1 dB input voltage of themain amplifier, the drive of the main amplifier towards compression, andhence towards increased efficiency, is not terminated prematurely inthis mode.

FIG. 5C is a plot of a gain curve 520 and an efficiency curve 522 of aDoherty-type amplifier operating with asymmetric post-transitionthreshold power division similar to the asymmetric post-transitionthreshold power division illustrated in FIG. 10A, in which thetransition threshold is selected to be substantially equal to the P1 dBinput voltage of the main amplifier of the Doherty-type amplifier. Ascan be seen from FIG. 5C, when the transition threshold is set to avalue that is substantially equal to the P1 dB input voltage of the mainamplifier, the main amplifier of the Doherty-type amplifier is driven toa level such that maximum efficiency is achieved. If the majority of thesubsequent input drive signal to the Doherty-type amplifier isredirected to the auxiliary amplifier after the transition threshold,the main amplifier is not overdriven, which means that pre-distortionalgorithms could potentially more easily linearize a Doherty-typeamplifier operated in this mode and the gain curve 520 does not have thesteep drop-off generally indicated in the gain curve 510 illustrated inFIG. 5B.

Reference is now made to FIGS. 6A to 6C in order to explain why it maybe advantageous to drive at least some of the subsequent input drivesignal of the Doherty-type amplifier to the main amplifier above thetransition threshold voltage.

FIGS. 6A and 6B are schematic diagrams of a Doherty-type amplifier 610that illustrate the impedances “seen” by the main and auxiliaryamplifiers when the auxiliary amplifier is turned on (FIG. 6A) andturned off (FIG. 6B). The Doherty-type amplifier 610 illustrated inFIGS. 6A and 6B includes a main amplifier 600, and auxiliary amplifier602, a combining structure 604 that combines outputs of the main andauxiliary amplifiers, and a matching structure 605 that matches thecombined output to a load 608. In FIGS. 6A and 6B, the load 608 is a 50Ohm load and the output matching structure 605 converts the 50 Ohm loadto be seen as a 25 Ohm load at the output of the combining structure604.

In FIGS. 6A and 6B, the Doherty output combining structure 604 isimplemented with a simple quarter wave transformer. If the auxiliaryamplifier 602 is biased in Class B, as described herein, then it willpresent an open circuit, i.e., high impedance, to the Doherty outputcombiner structure 604 when no signal is applied to the input of theauxiliary amplifier. This mode is illustrated in FIG. 6A, in which theauxiliary amplifier 602 presents an effective open circuit to the outputof the combining structure 604. The Doherty output combining structure604 is implemented with a simple quarter wave transformer, which in themode of operation shown in FIG. 6A transforms the 25 Ohm outputimpedance of the combining structure such that 100 Ohm is presented tothe main amplifier output.

However, with the auxiliary amplifier 602 biased in class B, as soon asany signal is applied to the auxiliary amplifier input the auxiliaryamplifier is turned on and the open circuit condition is removed, whichmeans that 50 Ohm is presented to both the main and auxiliary amplifieroutputs and the quarter-wave transformer 604 simply ensures that themain and auxiliary amplifier output signals combine in phase.Effectively, this means that once some portion of an input drive signalof the Doherty-type amplifier 610 is driven to the auxiliary amplifier602, for example, at the transition threshold described herein, and theauxiliary amplifier no longer presents an open circuit at its output,the output impedance presented to the main amplifier 600 drops from 100Ohm pre-transition to 50 Ohm post-transition.

FIG. 6C is a plot of two pairs of gain curves 610 and 614 and efficiencycurves 612 and 616 for an amplifier, such as the main amplifier 600shown in FIGS. 6A and 6B, presented with 100 Ohm and 50 Ohm loadconditions, respectively. It should be noted that the P1 dB inputvoltage of the amplifier under a 50 Ohm load condition is approximately3 dB greater than when terminated with 100 Ohm. This can clearly be seenin FIG. 6C, as the gain curve 610 drops off, i.e., compresses, at anoutput voltage that is approximately 3 dB less than the gain curve 614and the efficiency curve 616 has a peak at an output power that isapproximately 3 dB higher than the output power at which the efficiencycurve 612 peaks.

Accordingly, since the P1 dB compression point of the main amplifier ina Doherty-type amplifier will increase by approximately 3 dB after someinput signal is redirected to the auxiliary amplifier at the transitionthreshold, provided the auxiliary amplifier is biased in class B mode,it may be advantageous to continue to increase the input power of themain amplifier after transition, so that the main amplifier is driven toits “new” P1 dB compression point after the transition threshold.

Further discussion of potential post-transition power divisionstrategies for embodiments in which the transition threshold voltage istargeted as the P1 dB compression input voltage of the main amplifier isnow provided with reference to FIGS. 7A to 7C, which are plots of gainand efficiency versus output power for three post-transition powerdivision strategies in a Doherty-type amplifier in accordance withembodiments of the present invention.

FIG. 7A is a plot of gain 700 and efficiency 702 versus output power foran embodiment in which all input subsequent input power to thetransition threshold voltage is directed to the auxiliary amplifier,which is biased in class B. Here the reduction in output power providedby the main amplifier, generally indicated at 704, is evident as theoutput power does not reach 55 dBm and the associated efficiency isreduced.

FIG. 7B is a plot of gain 710 and efficiency 712 versus output power foran embodiment in which all subsequent input above the transitionthreshold voltage is split equally between the main and auxiliaryamplifier. Here peak power and efficiency issues present in the resultsshown in FIG. 7A are resolved, as full potential of the main amplifierwhen terminated with 50 Ohm post-transition is achieved, as generallyindicated at 714. However, the signal level to the main amplifier is toohigh as the device is now overly compressed beyond its P1 dB targetlevel, which can potentially lead to difficulties in linearizing theamplifier.

FIG. 7C is a plot of gain 720 and efficiency 722 versus output power foran embodiment in which all subsequent input above the transitionthreshold is split asymmetrically between the main and auxiliaryamplifiers. Here, as in FIG. 7B, the maximum power and efficiency issuesare resolved, but unlike the results shown in FIG. 7B the gain 720 doesnot compress much beyond 1 dB until it nears the max power level.

In some embodiments, a ratio between the portion of the subsequent inputabove the transition threshold voltage that is directed to the mainamplifier and the portion of the subsequent input above the transitionthreshold voltage that is directed to the auxiliary amplifier is basedupon the maximum power ratings of the main and auxiliary amplifier.

The performance impacts of the electrical length and bandwidth of thecombining and matching structures utilized in Doherty-type amplifiersare now discussed with reference to FIGS. 11 to 17.

The electrical length and the bandwidth of the matching structures usedto match the main and auxiliary amplifiers in a Doherty-type amplifierimpact the overall performance of the Doherty-type amplifier in back-offand through to saturation of the main amplifier. Specifically, thebandwidth and electrical length of the matching structures impact thetarget Cripps impedance presented to the main and auxiliary amplifiers,i.e., how well does the match correspond to the ideal Cripps impedanceacross frequency, and also impact the “off” impedance of the auxiliaryamplifier that is transformed to the Doherty combiner structure, whichwould ideally be an open-circuit, i.e. infinite impedance. The 1 dBcompression input voltage, gain and efficiency of the Doherty-typeamplifier across a given frequency band of operation may all be effectedby the electrical length and bandwidth of the matching structuresutilized in the Doherty-type amplifier.

It should be noted that electrically “short” matching structures aregenerally narrow band and broader band matching structures generally addsome additional electrical length in order to be realized.

The electrical length and bandwidth of the Doherty combining structureused to combine outputs of the main and auxiliary amplifiers in aDoherty-type amplifier only impact the performance of the Doherty-typeamplifier in back-off. Specifically, the bandwidth of the impedancepresented to the main amplifier in back-off is effected by the Dohertycombining structure. In general, narrow band designs will introduceimaginary components in the impedance presented to the main amplifier inback-off, which can negatively impact performance, as the idealimpedance to be presented to the main amplifier in back-off is generallya purely real impedance, as described below with reference to FIGS. 11Aand 11B.

The fundamental structure within a Doherty combining structure is theadditional quarter wavelength length included in the main amplifierpath. This length maintains quadrature between the main and auxiliaryamplifier paths when both devices are on and acts as an impedancetransformer when the auxiliary amplifier is off, as described hereinwith reference to FIGS. 6A and 6B. Typically, the additional quarterwavelength length in the main amplifier path is realized with a quarterwavelength length of microstrip with an impedance of X ohms, where X isan intermediate impedance of the design. For example, in FIGS. 6A and6B, the quarter wavelength microstrip 604 has an impedance of 50 Ohms.

FIGS. 13A and 13B are schematic diagrams of Doherty-type amplifiers 1300and 1310, respectively, with two alternative structures for realizingthe additional quarter wavelength length in the main amplifier path ofthe Doherty-type amplifier.

In FIG. 13A, the Doherty-type amplifier 1300 includes a main amplifier1302 and an auxiliary amplifier 1304 with a quarter wavelengthmicrostrip 1306 included in the main amplifier path in order to realizethe additional quarter wavelength length.

In FIG. 13B, the Doherty-type amplifier 1310 includes a main amplifier1312, an auxiliary amplifier 1314 and a more complex structure, whichincludes a shorted stub 1316, is used to implement the additionalquarter wavelength length in the main amplifier path.

FIGS. 14A and 14B are plots of simulated results for the s-parameter s11and the phase of the s-parameter s21, respectively, across frequency forthe quarter wave microstrip 1306 and the shorted stub 1316 shown inFIGS. 13A and 13B. The s-parameter s11 is an indication of how well theinput impedance presented by an element matches a target impedance,commonly 50 Ohm, while the phase of the s-parameter s21 indicates thephase transformation that the element imparts to a signal passingthrough it. Ideally, the element utilized to implement the additionalquarter wavelength length in the main amplifier path would present thetarget impedance, i.e., that impedance which is ideally presented to themain amplifier when both amplifiers are on, for all frequencies ofoperation and the element would also present a quarter wavelengthtransformation, i.e., 90 degree phase shift, across all frequencies ofoperation.

The simulated s-parameter s11 results 1400 and 1402 shown in FIG. 14Afor the quarter wavelength microstrip 1306 and the shorted stub 1316,respectively, clearly illustrate that for operating frequencies around2.1 GHz a typical 30 dB bandwidth for a quarter wavelength microstripprovides only a narrowband match on the order of 110 MHz, while ashorted stub provides a broader match with a 30 dB bandwidth ofapproximately 420 MHz.

However, while the shorted stub 1316 may offer a broader match atoperating frequencies around 2.1 GHz compared to the simple quarterwavelength microstrip 1306, the phase transformation provided by theshorted stub 1316, and other similar complex quarter wavelengthstructures, are typically less easily controlled across frequency.

The phase curve 1412 of the simulated s-parameter s21 for the shortedstub 1316 compared to the phase curve 1410 for the simple quarterwavelength microstrip 1306 illustrated in FIG. 14B clearly illustratethis design trade-off as the phase curve 1412 falls off more rapidlythan the phase curve 1410 and also changes from leading to lagging phaseat approximately 2.6 GHz. This rapid change from leading to laggingphase may not be a problem if it occurs outside of the frequency rangeof operation. For example, if the frequency range is assumed to be the30 dB bandwidth of the shorted stub, the change in the phase curve 1412at 2.6 GHz is outside the 30 dB bandwidth of the shorted stub indicatedby the simulated s11 curve 1402 shown in FIG. 14A.

It should be appreciated that, in general, any quarter wavelengthstructure used to implement the additional quarter wavelength length inthe main amplifier path of a Doherty-type amplifier that maintains boththe integrity of the signal quadrature and the desired impedancetransformation, while also improving overall bandwidth, will improve theperformance of the Doherty-type amplifier.

Stated briefly, the general design guidelines for the output structure,i.e., the matching structure associated with the main and auxiliaryamplifiers and the Doherty combining structure, of a Doherty-typeamplifier in accordance with an embodiment of the present inventioninclude:

i) the electrical length of the auxiliary amplifier path is equal ton*(λ/2), or n*180°, where n is an integer;

ii) the electrical length of the main amplifier path is ¼, or 90°,longer than that of the auxiliary amplifier;

iii) broadband design practices/techniques are to be used if possible;and

iv) the electrical length of the output structure is kept as short aspossible.

In general, point iii) is often in conflict with the other guidelines,as broadband design practices/techniques, impedance matching andtransformations often introduce additional electrical length. It hasbeen found that one way to overcome this conflict is to incorporate theDoherty combining structure and as much of the matching structure aspossible at the device, or even at the die, level, as any connectionsnecessary to interconnect devices or dies to external matching orcombining structures results in additional electrical length.

This point will now be demonstrated by way of example with reference toFIGS. 11, 12 and 15 to 17.

FIGS. 11A is a schematic diagram of an “ideal” Doherty combiningstructure 1100 that includes a first ideal quarter wavelength microstrip1102 in the main amplifier path and a second ideal quarter wavelengthmicrostrip 1104 at outputs of the main amplifier path and the auxiliaryamplifier path. The Doherty combining structure 1100 is considered“ideal”, since it is of minimum electrical length and maintains theintegrity of the impedance transformation and quadrature of the main andauxiliary amplifier signals.

FIG. 11B is a smith chart plot of the simulated s-parameter s22 of theDoherty combiner structure 100 shown in FIG. 11A.

The smith chart plot shown in FIG. 11B represents the simulateds-parameter s22 across the frequency range of 1.28 GHz to 3 GHz. Thes-parameter s22 is indicative of how well the output impedance of theDoherty combiner structure 1100 matches a desired load impedance. Threeresults 1100, 1112 and 1114 for operating frequencies of 2.11 GHz, 2.14GHz and 2.17 GHz are noted in FIG. 11B. These three results 1100, 1112and 1114 are grouped to the right hand side of the smith chart near thehorizontal axis of the Smith chart, which suggests that a broadbandstructure providing “purely” real impedance values may be an ideal “realworld” implementation of a Doherty combining structure.

FIG. 12A is a schematic diagram of a more realistic implementation 1200of a Doherty combiner structure than the “ideal” implementation 1100shown in FIG. 11A. The implementation 1200 shown in FIG. 12 includesgeometric elements representing examples of the physical components thatmay be used to implement and connect the ideal microstrips 1102 and 1104shown in FIG. 11A.

FIG. 12B is a Smith chart plot of the simulated s-parameter s22 of theDoherty combiner structure 1200 shown in FIG. 12B. As in FIG. 11B, thesmith chart plot shown in FIG. 12B represents the simulated s-parameters22 across the frequency range of 1.28 GHz to 3 GHz. In FIG. 12B, thesimulated result for s22 clearly shows that the improvisedimplementation shown in FIG. 12A introduces several wavelengths ofelectrical length, which emphasizes any error in the presentedimpedances. This is clearly illustrated by the s22 results 1210, 1212and 1214 for frequencies of 2.11 GHz, 2.14 GHz and 2.17 GHz,respectively, which have deviated from the horizontal axis of the Smithchart and spread out in phase. This means that even if the s22 curve isrotated back such that the s22 result 1212 at 2.14 GHz is located on thehorizontal axis of the Smith chart, i.e. purely real impedance, thespread in phase between the results 1210 and 1214 representing positiveand negative reactances would effectively detune the load reducing theperformance of the amplifier.

The simulation results in FIGS. 11A and 12A do not include the length ofthe associated device matching structure, e.g. the matching structure atthe output of the Doherty-type amplifier after the Doherty combinerstructure. Since the additional electrical length of any associateddevice matching structure would increase the total electrical length ofthe output structure, and would therefore emphasize any error in thepresented impedances, the electrical lengths of Doherty amplifiermatching structures are kept as short as possible in some embodiments ofthe present invention.

However, as noted above, electrically short structures are typicallynarrowband structures, which leads to a condition of diminishing returnsas the electrical length of the matching structure is reduced, sinceeventually the bandwidth of the match may become so reduced as aconsequence that it has a greater impact than the bandwidth of thecombining structure.

In order to further illustrate the role that bandwidth and electricallength of the output structure play in the performance of a Doherty-typeamplifier, a Doherty-type amplifier utilizing two Infineon GM8 poweramplifiers as the main and auxiliary amplifier was constructed and theperformance of the constructed Doherty-type amplifier was tested acrossfrequency. A schematic diagram of the test setup 1600 of theDoherty-type amplifier is shown in FIG. 16. The test setup includes aninput signal splitter 1604, a main amplifier 1606 (Infineon GM8) anauxiliary amplifier 1608 (Infineon GM8), a Doherty combiner structure1614 and one or more SMA connectors 1610 at the output of the Dohertycombiner structure 1614 to provide connection to measurement equipmentused to measure the s-parameter s22 at the output of the Dohertycombiner structure.

FIG. 15A is a Smith chart plot of the measured s-parameter s22 1505 forthe test setup 1600 shown in FIG. 16, with the measurement equipmentcalibrated to a reference plane 1602A between the output of the Dohertycombiner structure 1614 and the input of the SMA connector 1610 shown inFIG. 16. Calibration at the reference plane 1602A effectively removesthe effect of the electrical length of the SMA connector 1610 from themeasured results. The measured s-parameter s22 at three frequencies,2.11 GHz, 2.14 GHz and 2.17 GHz, are highlighted at 1500, 1502 and 1504,respectively, in FIG. 15A. These results represent a narrowband solutionof “purely” real values at 2.14 GHz, as the results 1502 for 2.14 GHz istoward the right hand edge and near the horizontal “real” axis of theSmith chart, while the results 1500 and 1504 at 2.11 GHz and 2.17 GHz,respectively, contain significant reactance components.

In order to demonstrate the performance impact of the electrical lengthof the Doherty output structure, which typically includes a Dohertycombiner, such as the Doherty combiner structure 1614 shown in FIG. 16,and any associated matching structure, the measurement equipment wasrecalibrated to a second reference plane 1602B, as shown in FIG. 16.This recalibration allowed the effect of the electrical length of theSMA connector 1610 to be included in the measured s-parameter s22.

FIG. 15B is a Smith chart plot of the measured s-parameter s22 1515 forthe test setup 1600 shown in FIG. 16, with the measurement equipmentcalibrated to the reference plane 1602B. As in FIG. 15A, the measureds-parameter s22 at 2.11 GHz, 2.14 GHz and 2.17 GHz, are highlighted at1510, 1512 and 1514, respectively, in FIG. 15B. It is clear from themeasured results shown in FIG. 15B that the electrical length of the SMAconnector 1610, which is now included the measured results, has causedthe measured s-parameter s22 to rotate away from the “purely” realnarrowband results indicated at 2.14 GHz in FIG. 15A for the calibrationreference plane 1602A.

By varying the electrical length of the SMA connection between theDoherty combiner structure 1614 and the measurement equipment by usingone or more SMA connectors with a different total combined electricallength different than the electrical length of the SMA connector thatproduced the results 1515 shown in FIG. 15B, it was found that it waspossible to “tune” the SMA interconnection with a different combinedelectrical length of the SMA connection for each of the threefrequencies 2.11 GHz, 2.14 GHz and 2.17 GHz. FIG. 15A demonstrates thata “purely” real result was obtained at 2.14 GHz with the electricallength of the original SMA connector 1610. In order to mitigate thepositive and negative reactances in the results 1500 and 1504 at 2.11GHz and 2.17 GHz, respectively, measurements of the s-parameter s22 wererepeated first with an SMA connector with a longer overall electricallength than the original SMA connector 1610 and then again with an SMAconnector with a shorter overall electrical length than the original SMAconnector.

FIG. 15C is a Smith chart plot of the measured s-parameter s22 with theoriginal SMA connector 1610, generally indicated at 1515, with theshorter SMA connector, generally indicated at 1526, and with the longerSMA connector, generally indicated at 1530. The results at 2.11 GHz,2.14 GHz and 2.17 GHz for the three curves 1530, 1515 and 1526,respectively, are highlighted at 1520, 1512 and 1524 in FIG. 15C. As canbe seen in FIG. 15C, the different electrical lengths of the SMAconnector at the output of the Doherty-type amplifier for the threeconfiguration measurements 1530, 1515 and 1526 shown in FIG. 15C hascaused the s-parameter s22 1520 measured at 2.11 GHz for the longerconfiguration and the s-parameter s22 1524 measured at 2.17 GHz for thelonger configuration to converge with the s-parameter s22 1515 measuredfor the original configuration. This convergence demonstrates that thedetuning caused by the electrical length of the original SMA connector,i.e., the reactances in the measured s-parameter s22 results 1500 and1504 at 2.11 GHz and 2.17 GHz shown in FIG. 15A, could be mitigated byreducing the overall electrical length of the SMA connector 1610.

It should be noted that it is assumed that the convergence in theresults shown in FIG. 15C, which were made with the measurementcalibrated to the second reference plane 1602B, will translate back downto the initial reference plane 1602A at the output of the Dohertycombiner structure 1614.

The performance impacts of the s22 optimization illustrated in FIG. 15Care now discussed with reference to FIGS. 17A to 17C.

FIG. 17A is a plot of measured gain versus output power across frequencyfor the three SMA connector configurations described above withreference to FIGS. 15C and 16. FIG. 17A includes three results 1700,1702 and 1704 for the gain of the Doherty-type amplifier for theoriginal SMA connector at frequencies of 2.11 GHz, 2.14 GHz and 2.17GHz, respectively. FIG. 17A also includes three gain curves 1706, 1708and 1710 for the longer SMA connector, the original SMA connector, andthe shorter SMA connector, respectively, at frequencies of 2.11 GHz,2.14 GHz and 2.17 GHz, respectively. It is noted that the curves 1702and 1708 are both measurements made at 2.14 GHz with the original SMAconnector, and therefore should agree with one another as shown in FIG.17A.

The results in FIG. 17A clearly show a more consistent gain profile inback-off across frequencies for the optimized configurations.Specifically, the gain profiles 1706, 1708 and 1710 have converged inthe region generally indicated at 1712 in FIG. 17A. However, theremaining variance in the gain profiles above 45 dBm output powersuggests that further optimization is possible.

FIG. 17B is a plot of efficiency versus output power across frequencyfor the three SMA connector configurations described above withreference to FIGS. 15C and 16. FIG. 17B includes three results 1714,1716 and 1718 for the efficiency of the Doherty-type amplifier for theoriginal SMA connector at frequencies of 2.11 GHz, 2.14 GHz and 2.17GHz, respectively. FIG. 17B also includes three efficiency curves 1720,1722 and 1724 for the longer SMA connector, the original SMA connector,and the shorter SMA connector, respectively, at frequencies of 2.11 GHz,2.14 GHz and 2.17 GHz, respectively. It is noted that the efficiencycurves 1716 and 1722 are both measurements made at 2.14 GHz with theoriginal SMA connector, and therefore should agree with one another asshown in FIG. 17B.

In accordance with the improvement seen in the gain profiles seen in theback-off region of operation in FIG. 17A, the efficiency curves 1720,1722 and 1724 for the optimized configurations plotted in FIG. 17B showimproved consistency in the back-off region of operation below 47 dBmoutput power and the droop in back-off in the efficiency curve 1714 forthe original configuration operating at 2.11 GHz has been mitigated, asclearly indicated at 1728 in FIG. 17B.

For the sake of comparison, FIG. 17B also includes an efficiency curve1726 for a conventional Class AB biased power amplifier operating at2.14 GHz. The first peak in the efficiency curves 1722 for theDoherty-type amplifier provides an increase in efficiency of almost 13%,generally indicated at 1730 in FIG. 17B, over the efficiency curve 1726for the conventional Class AB biased power amplifier at 48 dBm of outputpower. Again, the variance in the efficiency curves 1720, 1722 and 1724in the saturation region above 48 dBm of output power suggests thatfurther optimization may be possible.

FIG. 17C is a plot of the phase introduced by the Doherty-type amplifierversus output power across frequency for the three SMA connectorconfigurations described above with reference to FIGS. 15C and 16. FIG.17C includes three results 1732, 1734 and 1736 for the phase changeintroduced by the Doherty-type amplifier with the original SMA connectorat frequencies of 2.11 GHz, 2.14 GHz and 2.17 GHz, respectively. FIG.17C also includes three phase curves 1738, 1740 and 1742 for the longerSMA connector, the original SMA connector, and the shorter SMAconnector, respectively, at frequencies of 2.11 GHz, 2.14 GHz and 2.17GHz, respectively.

It is noted in FIG. 17C that after optimization there is a generalconvergence of AM-PM (Amplitude Modulation to Phase Modulation) responseacross frequency in saturation, as evidenced by the convergence of thephase curves 1738, 1740 and 1742 in the region generally indicated at1744, whereas, the phase curves 1732, 1734 and 1736 for the originalconfiguration vary widely in that region. AM-PM response is an unwantedconversion of amplitude variation into phase variation in the output ofthe amplifier. Accordingly, the convergence in the AM-PM response acrossfrequency of the optimized configurations represents a performanceimprovement.

It should be understood that as used herein, terms such as coupled,connected, electrically connected, in signal communication, and the likemay include direct connections between components, indirect connectionsbetween components, or both, as would be apparent in the overall contextof a particular embodiment. The term coupled is intended to include, butnot be limited to, a direct electrical connection.

It must be further understood that the simulated and measured resultsdescribed herein and illustrated in the Figures are provided by way ofexample only and were made under conditions. Under other actual orsimulation conditions, similar or possibly different results may beachieved.

Although this disclosure describes specific implementations ofDoherty-type amplifiers using semiconductor devices with unequal, orasymmetric, power ratings, i.e., Asymmetric or Enhanced asymmetricDoherty-type amplifiers, the same or similar approach can be used forall Doherty-type architectures, including a traditional symmetricDoherty-type design, in order to achieve higher gain.

The foregoing description includes many detailed and specificembodiments that are provided by way of example only, and should not beconstrued as limiting the scope of the present invention. Alterations,modifications and variations may be effected to the particularembodiments by those of skill in the art without departing from thescope of the invention, which is defined solely by the claims appendedhereto.

1. An amplifier arrangement for amplifying an amplifier arrangementinput, the amplifier arrangement comprising: a main amplifier pathcomprising a main amplifier; at least one auxiliary amplifier pathcomprising an auxiliary amplifier; a combining structure configured tocombine outputs of the main and auxiliary amplifiers; a signalpreparation element operable to develop a main amplifier input signalfor coupling to the main amplifier path and at least one auxiliaryamplifier input signal for coupling to the at least one auxiliaryamplifier path as a function of the amplifier arrangement input and atransition threshold associated with the amplifier arrangement input;and a bias controller operable to bias the at least one auxiliaryamplifier as a class B amplifier.
 2. The amplifier arrangement of claim1, wherein the bias controller is operable to bias the auxiliaryamplifier substantially at a turn-on voltage of the auxiliary amplifier.3. The amplifier arrangement of claim 1, wherein the signal preparationelement is operable: to develop the main amplifier input signal withsubstantially all of the amplifier arrangement input for amplifierarrangement input levels that are below the transition threshold; and toredirect at least a portion of the amplifier arrangement input todevelop the at least one auxiliary amplifier input for amplifierarrangement input levels that are above the transition threshold.
 4. Theamplifier arrangement of claim 3, wherein the signal preparation elementis operable to asymmetrically divide the amplifier arrangement inputabove the transition threshold between the main amplifier input signaland the at least one auxiliary amplifier input signal.
 5. The amplifierarrangement of claim 4, wherein the signal preparation element isoperable to asymmetrically divide the amplifier arrangement input abovethe transition threshold based on a ratio between power ratings of themain amplifier and the at least one auxiliary amplifier.
 6. Theamplifier arrangement of claim 1, comprising a plurality of auxiliaryamplifier paths, wherein the signal preparation element is operable todevelop the main amplifier input signal for the main amplifier path andto develop a respective auxiliary amplifier input for each of theauxiliary amplifier paths based on a plurality of respective transitionthresholds.
 7. The amplifier arrangement of claim 1, further comprisinga matching structure operable to substantially match outputs of the mainamplifier path and the at least one auxiliary amplifier path to a loadimpedance.
 8. The amplifier arrangement of claim 1, wherein the mainamplifier and the at least one auxiliary amplifier are asymmetricallysized in terms of power ratings.
 9. The amplifier arrangement of claim1, wherein the main amplifier and the at least one auxiliary amplifierare fabricated in different semiconductor technologies.
 10. Theamplifier arrangement of claim 1, wherein the main amplifier and the atleast one auxiliary amplifier are in a common package.
 11. A method ofcontrolling an amplifier arrangement, the amplifier arrangementcomprising a main amplifier path and at least one auxiliary amplifierpath, the main amplifier path comprising a main amplifier and the atleast one auxiliary amplifier path comprising an auxiliary amplifier,the method comprising: biasing the at least one auxiliary amplifier as aclass B amplifier; and developing a main amplifier input signal forcoupling to the main amplifier path and developing at least oneauxiliary amplifier input signal for coupling to the at least oneauxiliary amplifier path as a function of the amplifier arrangementinput and a transition threshold associated with amplifier arrangementinput.
 12. The method of claim 11, comprising biasing the auxiliaryamplifier substantially at a turn-on voltage of the auxiliary amplifier.13. The method of claim 11, wherein developing the main amplifier inputsignal and the auxiliary amplifier input signal comprises: developingthe main amplifier input signal with substantially all of the amplifierarrangement input for amplifier arrangement input levels that are belowthe transition threshold; and redirecting at least a portion of theamplifier arrangement input to develop the at least one auxiliaryamplifier input for amplifier arrangement input levels that are abovethe transition threshold.
 14. The method of claim 13, wherein the signalpreparation element is operable to asymmetrically divide the amplifierarrangement input above the transition threshold between the mainamplifier input signal and the at least one auxiliary amplifier inputsignal.
 15. The method of claim 14, wherein redirecting at least aportion of the amplifier arrangement input to develop the at least oneauxiliary amplifier input for amplifier arrangement input levels thatare above the transition threshold comprises asymmetrically dividing theamplifier arrangement input above the transition threshold based on aratio between power ratings of the main amplifier and the at least oneauxiliary amplifier.
 16. The method of claim 11, wherein the amplifierarrangement comprises a plurality of auxiliary amplifier paths, whereindeveloping a main amplifier input signal for coupling to the mainamplifier path and developing at least one auxiliary amplifier inputsignal comprises developing the main amplifier input signal for the mainamplifier path and developing a respective auxiliary amplifier input foreach of the auxiliary amplifier paths based on a plurality of respectivetransition thresholds.
 17. The method of claim 11, further comprisingmatching outputs of the main amplifier path and the at least oneauxiliary amplifier path to a load impedance.
 18. The method of claim11, wherein the main amplifier and the at least one auxiliary amplifierare asymmetrically sized in terms of power ratings.
 19. The method ofclaim 11, wherein the main amplifier and the at least one auxiliaryamplifier are fabricated in different semiconductor technologies. 20.The method of claim 11, wherein the main amplifier and the at least oneauxiliary amplifier are in a common package.
 21. A base stationcomprising the amplifier arrangement of claim
 1. 22. A mobilecommunication device comprising the amplifier arrangement of claim 1.